Carrier restoration apparatus and method

ABSTRACT

A carrier restoration apparatus for acquiring a frequency offset and tracking a phase jitter from a pass-band digital signal having the frequency offset and the phase jitter is disclosed. In the apparatus, a frequency acquisition PLL section for acquiring the frequency offset and a phase tracking PLL section for tracking the residual phase jitter are separately constructed, and the apparatus operates in two modes for first acquiring the frequency offset and then tracking the residual phase jitter. Thus, a rapid acquisition/tracking is performed so as to minimize the frequency offset and phase jitter of several hundred KHz produced from a tuner or an RF oscillator, and a high-reliability acquisition/tracking can be performed even under the low SNR and serious channel ISI (i.e., ghost).

BACKGROUND OF THE INVENTION

[0001] 1. Field of the Invention

[0002] The present invention relates generally to a quadrature amplitudemodulation/phase shift keying (QAM/PSK) receiver, and more particularly,to a carrier restoration apparatus and method that compensates for afrequency offset and a phase jitter of a carrier.

[0003] 2. Background of the Related Art

[0004] Typically, a quadrature amplitude modulation (QAM) is used forcable transmission/reception of compressed digital video data in a HDTV.Especially, the 256 QAM modulation is performed in a manner that thecompressed video data is encoded for transmission to output 256constellations corresponding to 8 bits per symbol period (i.e., 5.3607MHz) as vector values, orthogonal projected values of the vector valueson orthogonal axes I and Q are carrier-suppression-modulated by sine andcosine waves, respectively, and then the modulated waves are addedtogether to be transmitted.

[0005] In order for a receiving end to obtain the vector values of the256 constellations again by demodulation, it is required to restore thecarrier that is phase-synchronized with the carrier of the receivedsignal and has not been modulated. That is because the orthogonalprojection values of the vector values of the 256 constellations on theorthogonal axes I and Q can be obtained by multiplying the receivedsignal by the sine and cosine waves phase-synchronized with the carrierof the received signal, respectively.

[0006] Specifically, a carrier restoration section mounted on the QAMreceiver in the HDTV cable transmission system should rapidly acquireand track a frequency offset Δω of several hundred KHz and a residualphase jitter Δθ generated from a tuner or RF oscillator to minimizethem. Also, the carrier restoration section should perform ahigh-reliability acquisition/tracking operation even under a lowsignal-to-noise ratio (SNR) and a severe channel ISI (i.e., ghost).

[0007]FIG. 1 is a block diagram illustrating the construction of ageneral television (TV) receiver. According to this TV receiver, apreprocessing section 11 outputs to a carrier restoration section 12 apass-band digital signal having a frequency offset and phase jitter. Thecarrier restoration section 12 modulates the pass-band digital signaloutputted from the preprocessing section 11 into sine/cosine waves togenerate a base-band digital signal from which the frequency offset andthe phase jitter are removed. The base-band digital signal is outputtedto a post-processing section 13.

[0008] For example, if it is assumed that the carrier restorationsection of FIG. 1 restores the carrier of the signal modulated by theQAM, the effect exerted by a phase error at that time is as follows.

[0009] That is, if it is defined that I(t) and Q(t) are inphase andquadrature base-band signals, and a modulated signal is fc, aQAM-modulated signal S(t) is expressed by the following equation 1.

S(t)=I(t)*cos(2πƒ_(c) t)−Q(t)*sin(2πƒ_(c) t)  [Equation 1]

[0010] If the modulated signal is then demodulated into two inphase andquadrature carrier waves having a phase error φ, the base-band signalsas shown in the following equations 2 and 3 are obtained.$\begin{matrix}\begin{matrix}{{{DI}(t)} = \quad {{LPF}\left\{ {{S(t)}*{\cos \left( {{2\pi \quad f_{c}t} + \varphi} \right)}} \right\}}} \\{= \quad {{\left( {1/2} \right)*{I(t)}*{\cos (\varphi)}} - {\left( {1/2} \right)*{Q(t)}*{\sin (\varphi)}}}}\end{matrix} & \left\lbrack {{Equation}\quad 2} \right\rbrack \\\begin{matrix}{{{DQ}(t)} = \quad {{LPF}\left\{ {{S(t)}*{\sin \left( {{2\pi \quad f_{c}t} + \varphi} \right)}} \right\}}} \\{= \quad {{\left( {1/2} \right)*{I(t)}*{\sin (\varphi)}} - {\left( {1/2} \right)*{Q(t)}*{\cos (\varphi)}}}}\end{matrix} & \left\lbrack {{Equation}\quad 3} \right\rbrack\end{matrix}$

[0011] cos(φ) of the first term of Equation 2 and of the second term ofEquation 3 represents the gain error, and sin(s) of the second term ofEquation 2 and of the third term of Equation 3 represents an errorcaused by interference.

[0012] As described above, in case of the QAM, the phase error of therestored carrier has an effect on not only the gain error but also theerror due to the interference, and thus this causes its effect to becomemore serious.

[0013] Accordingly, two conventional method of restoring the carrier inthe receiving end have been proposed to solve the above-describedproblem.

[0014] One of them is a method of extracting a pilot signal from thefrequency of a received signal, and synchronizing an output frequencyand phase of a local oscillator with those of the received signal in thereceiving end. This method is used for restoring the carrier of avestigial side band (VSB) that is the ground wave of the conventionalHDTV transmission system.

[0015] The other is a method of estimating the frequency and phase ofthe carrier directly from a suppression-modulated signal. This methodhas been widely used for the carrier restoration of the QAM and PSK ofthe conventional HDTV cable transmission system.

[0016] As the conventional carrier restoration method for estimating thefrequency and phase of the carrier directly from thesuppression-modulated signal, there have been proposed a square loopmethod as shown in FIG. 2, Costas loop method in FIG. 3, and decisionfeedback loop method in FIG. 4.

[0017] First, the square loop as shown in FIG. 2 restores the carrier ofa transmitted signal S(t) by modulating the signal by a double sideband/suppressed carrier (DSB/SC) phase amplitude modulation (PAM) asexpressed by the following equation 4.

S(t)=A(t)*cos(2πƒ_(c) t+φ)  [Equation 4]

[0018] In Equation 4, if the base-band signal level is symmetricalcentering around 0, the average expected value becomes 0 as shown in thefollowing equation 5.

E[S(t)]=E[A(t)]=0  [Equation 5]

[0019] Accordingly, any phase information cannot be obtained from theaverage value of the received signal. At this time, the square loop asshown in FIG. 2 may be used as a method of driving a phase locked loop(PLL) by extracting the frequency component from 2πƒ_(c)t.

[0020] Specifically, the output S²(t) of a square section 21 is obtainedby the following equation 6, and the average expected value is 0, thefrequency component can be extracted from 2πƒ_(c)t. $\begin{matrix}\begin{matrix}{\quad {{S^{2}(t)} = \quad {{A^{2}(t)}*{\cos^{2}\left( {{2\pi \quad f_{c}t} + \varphi} \right)}}}} \\{\left. {= \quad {{\left( {1/2} \right)*{A^{2}(t)}} + {1/2}}} \right)*{A^{2}(t)}*{\cos \left( {{4\pi \quad f_{c}t} + {2\varphi}} \right)}}\end{matrix} & \left\lbrack {{Equation}\quad 6} \right\rbrack\end{matrix}$

[0021] Accordingly, if the output S²(t) of the square section 21 passesthrough a band pass filter 22 having a center frequency of 2πƒ_(c), theDC component is removed, and only a component having a frequency of 2fc,phase of 2φ, and amplitude of ½* A²(t)*H(2fc) remains. Here, H(2fc) isthe gain of the band pass filter.

[0022] In order to synchronize the oscillated frequency of a localoscillator 25 with the output of a band pass filter 22, a PLL process isperformed. Specifically, the output of the band pass filter 22 and theoutput of the local oscillator 25 are multiplied through a multiplier23, and the multiplied output is inputted to a loop filter 24. Theoutput of the loop filter 24 is inputted to the local oscillator 25again. That is, the loop filter 24 filters and accumulates the output ofthe multiplier 23 to detect a phase error, and output the phase error tothe local oscillator 25. The local oscillator 25 generates a frequencysin(4πƒ_(c)t+2φ) that is in proportion to the phase error, and outputsthe generated frequency to the multiplier 23 and a frequency divider 26.

[0023] The frequency divider 26 divides the output of the localoscillator 25 to obtain a restored carrier of sin(4πƒ_(c)t+2φ). Here, θis an estimated value of φ, and the PLL is formed so as to effect φ−θ=0.

[0024] However, the carrier restoration by the above-described squareloop has a phase ambiguity of 180° with respect to the phase of thereceived signal since the local oscillator 25 is synchronized with thefrequency component of 2fc, and the restored carrier is generatedthrough the frequency divider 26. This problem can be solved in a mannerthat the transmitting end performs a differential encoding, and thereceiving end performs a differential decoding, but the frequencyambiguity still increases. Specifically, in case that the modulatedsignal contains information with M phases (i.e., the transmitted signalis given by the following equation 7), the frequency ambiguity increasesto 2π/M if an M-involution element is used in replace of the squareelement and the frequency divider performs % M.

S(t)=A(t)*cos[2πƒ_(c) t+φ+(2π/M)*(m−1)]where, m=1, 2, 3, . . .M.  [Equation 7]

[0025] Next, the Costas loop method will be explained.

[0026] The Costas loop as shown in FIG. 3 restores the carrier of thetransmitted signal expressed by Equation 4.

[0027] In FIG. 3, outputs Y_(c)(t) and Y_(s)(t) of first and secondmultipliers 31 and 32 can be expressed by the following equations 8 and9. $\begin{matrix}\begin{matrix}{{Y_{c}(t)} = \quad {\left\lbrack {{S(t)} + {N(t)}} \right\rbrack*{\cos \left( {{2\pi \quad f_{c}t} + \theta} \right)}}} \\{= \quad {{\left( {1/2} \right)*\left\lbrack {{A(t)} + {N_{c}(t)}} \right\rbrack*\cos \quad {\Delta\varphi}} +}} \\{\quad {{\left( {1/2} \right)*{N_{s}(t)}*\sin \quad {\Delta\varphi}} + {2f_{c}}}}\end{matrix} & \left\lbrack {{Equation}\quad 8} \right\rbrack \\\begin{matrix}{\quad {{Y_{c}(t)} = \quad {\left\lbrack {{S(t)} + {N(t)}} \right\rbrack*{\sin \left( {{2\pi \quad f_{c}t} + \theta} \right)}}}} \\{= \quad {{\left( {1/2} \right)*\left\lbrack {{A(t)} + {N_{c}(t)}} \right\rbrack*\sin \quad {\Delta\varphi}} +_{c}}} \\{\quad {{\left( {1/2} \right)*{N_{s}(t)}*\cos \quad {\Delta\varphi}} + {2f}}}\end{matrix} & \left\lbrack {{Equation}\quad 9} \right\rbrack\end{matrix}$

[0028] Here, the components of Δφ=φ−θ, and 2f_(c) are removed passingthrough first and second base-band pass filters 32 and 36. The outputsof the first and second base-band pass filters 32 and 36 are multipliedby a multiplier 33 to produce an error signal as expressed by thefollowing equation 10. $\begin{matrix}\begin{matrix}{{e(t)} = \quad {{\left( {1/8} \right)*\left\{ {\left\lbrack {{A(t)} + {N_{c}(t)}} \right\rbrack^{2} - {N_{s}^{2}(t)}} \right\}*\sin \quad 2\quad \Delta \quad \varphi} -}} \\{\quad {\left( {1/4} \right)*{N_{s}(t)}*\left\lbrack {{A(t)} + {N_{c}(t)}} \right\rbrack*\cos \quad 2\quad \Delta \quad \varphi}}\end{matrix} & \text{[Equation~~10]}\end{matrix}$

[0029] In Equation 10, it can be recognized that the error signal e(t)is composed of a desired signal component of A²(t)*sin2Δφ, component ofsignal*noise, and component of noise*noise. Here, a matched filter maybe suitably used as the first and second base-band pass filters 32 and36. If the matched filter is used, the noise mixed to the loop can bereduced.

[0030] The operation of a loop filter 37 that received the output of themultiplier 33 and the operation of a local oscillator 28 are the same asthose in the above-described square loop method, and the detailedexplanation thereof will be omitted. That is, the Costas loop method isequivalent to the square loop method, and has the phase ambiguity of180° .

[0031] Next, the decision feedback loop method will be explained.

[0032] The above-described Costas loop method has the problem in that asthe error signal is multiplied by the noise, the noise is amplified toits square value. This problem can be solved by adding a decisionelement to one side of the Costas loop as shown in FIG. 3. This type ofcarrier restoration is called the decision feedback loop method, whichis illustrated in FIG. 4. Referring to FIG. 4, a sampler 43 and adecision element 45 are arranged between a first base-band pass filter42 and a multiplier 49 of the carrier restoration apparatus of FIG. 3.Here, the sampler 43 receives from a timing restoration section 44timing errors of present symbols produced through the base-band signalprocess, and performs an interpolation to reduce the errors among theoutput signals of the first base-band pass filter 42. Also, the decisionelement 45 generates and outputs to the multiplier 49 decision signalsmatching respective signal levels of the base-band signals outputtedfrom the sampler 43.

[0033] If there is no error in the decision element in FIG. 4, theoutput of the decision element 45 will be the base-band signal A(t) fromwhich the noise is removed. Accordingly, if the phase error signal e(t)is developed, the square component of the noise is vanished as shown inthe following equation 1. $\begin{matrix}\begin{matrix}{{e(t)} = \quad {\left( {1/2} \right)*{A(t)}*\left\{ {{\left\lbrack {{A(t)} + {N_{c}(t)}} \right\rbrack*\sin \quad \Delta \quad \varphi} -} \right.}} \\{\left. \quad {{N_{s}(t)}*\cos \quad {\Delta\varphi}} \right\} + {2f_{c}}} \\{= \quad {{\left( {1/2} \right)*{A^{2}(t)}*\sin \quad {\Delta\varphi}} + {\left( {1/2} \right)*}}} \\{\quad {{{A(t)}*\left\lbrack {{{N_{c}(t)}*\sin \quad {\Delta\varphi}} - {{N_{s}(t)}*\cos \quad {\Delta\varphi}}} \right\rbrack} + {2f_{c}}}}\end{matrix} & \text{[Equation~~11]}\end{matrix}$

[0034] However, the decision feedback loop method as described abovealso has the following problems.

[0035] First, since an elaborate high-quality tuner should be usedaccording to a small acquisition/tracking range, the cost for preparingthe tuner is increased. That is, a tuner with a small frequency offsetand small phase jitter during the carrier restoration has a goodperformance, and such a tuner having the good performance is typicallyexpensive.

[0036] Second, the BER performance of the receiver is lowered due to alarge residual phase jitter.

[0037] Third, the acquisition/tracking performance with respect to asmall input SNR deteriorates. That is because if the receiving power(i.e., SNR) of the input signal is small, the error detection section ofthe conventional carrier restoration section produces an inaccurateerror.

[0038] Fourth, the acquisition/tracking performance with respect to theISI/ghost channel deteriorates widely. Even in the channel having astrong ISI/ghost, the error detection section also produces aninaccurate error in the same manner.

SUMMARY OF THE INVENTION

[0039] Accordingly, the present invention is directed to a carrierrestoration apparatus and method that substantially obviates one or moreproblems due to limitations and disadvantages of the related art.

[0040] An object of the present invention is to provide a carrierrestoration apparatus and method which can improve the frequencyacquisition performance and the phase tracking performance by separatelyconstructing a loop for frequency acquisition and a loop for phasetracking.

[0041] Additional advantages, objects, and features of the inventionwill be set forth in part in the description which follows and in partwill become apparent to those having ordinary skill in the art uponexamination of the following or may be learned from practice of theinvention. The objectives and other advantages of the invention may berealized and attained by the structure particularly pointed out in thewritten description and claims hereof as well as the appended drawings.

[0042] To achieve these objects and other advantages and in accordancewith the purpose of the invention, as embodied and broadly describedherein, a carrier restoration apparatus comprises a phase/frequencydetection section for obtaining a phase error between demodulated signalconstellations and blind decision signal constellations ordecision-directed decision signal constellations, and extracting apolarity of the phase error, a PLL section for frequency acquisition forextracting a corresponding frequency offset by accumulatingpre-calculated bandwidth values according to the polarity of the phaseerror, generating digital type sine and cosine waves according to theextracted frequency offset, and then generating a base-band digitalsignal where the frequency offset of the carrier is acquired bydemodulating a pass-band digital signal by the sine and cosine waves, aPLL section for phase tracking for extracting a corresponding phasejitter by accumulating the pre-calculated bandwidth values according tothe polarity of the phase error, generating digital type sine and cosinewaves according to the extracted phase jitter, and then generating thedemodulated signal constellations where the phase jitter is tracked bydemodulating the base-band digital signal by the sine and cosine waves,a blind decision section for extracting the polarity of the demodulatedsignal constellations generated from the PLL section for phase tracking,and generating blind signal constellations by slicing the demodulatedsignal constellations according to the extracted polarity, and adecision-directed decision section for generating decision-directeddecision signal constellations matching respective signal levels of thedemodulated signal constellations generated from the PLL section forphase tracking.

[0043] It is preferable that the phase/frequency detection sectionoperates in a blind mode for extracting the polarity by obtaining thephase error between the demodulated signal constellation and the blinddecision signal constellations in order to acquire the frequency offset,or in a decision-directed mode for extracting the polarity by obtainingthe phase error between the demodulated signal constellation and thedecision-directed decision signal constellations in order to track thephase jitter. Thus, it is also preferable that the apparatus furthercomprises a lock detection section for controlling selection of theblind mode and the decision-directed mode of the phase/frequencydetection section.

[0044] Preferably, a carrier restoration method according to the presentinvention performs the above-described carrier restoration process bysoftware.

[0045] It is to be understood that both the foregoing generaldescription and the following detailed description of the presentinvention are exemplary and explanatory and are intended to providefurther explanation of the invention as claimed.

BRIEF DESCRIPTION OF THE DRAWINGS

[0046] The accompanying drawings, which are included to provide afurther understanding of the invention and are incorporated in andconstitute a part of this application, illustrate embodiment(s) of theinvention and together with the description serve to explain theprinciple of the invention. In the drawings:

[0047]FIG. 1 is a schematic view illustrating the construction of ageneral TV receiver;

[0048]FIG. 2 is a block diagram illustrating the construction of aconventional carrier restoration apparatus using a square loop method;

[0049]FIG. 3 is a block diagram illustrating the construction of aconventional carrier restoration apparatus using a Costas loop method;

[0050]FIG. 4 is a block diagram illustrating the construction of aconventional carrier restoration apparatus using a decision feedbackloop method;

[0051]FIG. 5 is a block diagram illustrating the construction of acarrier restoration apparatus according to the present invention appliedto a TV receiver;

[0052]FIG. 6 is a block diagram illustrating the detailed constructionof a phase/frequency detection section of FIG. 5;

[0053]FIG. 7 is a block diagram illustrating the detailed constructionof a blind decision section of FIG. 5;

[0054]FIG. 8 is a view illustrating an example of decision signalconstellations of a blind decision element of 4/16/64/256 QAM;

[0055]FIG. 9 is a block diagram illustrating the detailed constructionof a decision-directed decision element of FIG. 5;

[0056]FIG. 10 is a view illustrating an example of constellations of a16 QAM decision-directed decision signal;

[0057]FIG. 11 is a block diagram illustrating the detailed constructionof a frequency acquisition loop filter of FIG. 5;

[0058]FIG. 12 is a block diagram illustrating the detailed constructionof a numerically controlled oscillator (NCO) of FIG. 5;

[0059]FIG. 13 is a block diagram illustrating the detailed constructionof a frequency acquisition element of FIG. 5;

[0060]FIG. 14 is a block diagram illustrating the detailed constructionof a phase tracking loop filter of FIG. 5;

[0061]FIG. 15 is a block diagram illustrating the detailed constructionof a phase ROM table of FIG. 5;

[0062]FIG. 16 is a block diagram illustrating the detailed constructionof a phase tracking element of FIG. 5;

[0063]FIGS. 17A and 17B are views illustrating the geometricalcharacteristic of a characteristic function of a phase/frequencydetector in a blind mode, wherein FIG. 17A shows an example in case thatthe phase of the demodulated signal constellations is larger than thephase of the decision signal constellations, and FIG. 17B shows anexample in case that the phase of the demodulated signal constellationsis smaller than the phase of the decision signal constellations of thedemodulated signal; and

[0064]FIGS. 18A and 18B are views illustrating the geometricalcharacteristic of a characteristic function of a phase/frequencydetector in a decision-directed mode, wherein

[0065]FIG. 18A shows an example in case that the phase of thedemodulated signal constellations is larger than the phase of thedecision signal constellations, and FIG. 18B shows an example in casethat the phase of the demodulated signal constellations is smaller thanthe phase of the decision signal constellations of the demodulatedsignal.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0066] Reference will now be made in detail to the preferred embodimentsof the present invention, examples of which are illustrated in theaccompanying drawings.

[0067]FIG. 5 is a block diagram illustrating the construction of acarrier restoration apparatus according to the present invention appliedto a TV receiver. Referring to FIG. 5, a carrier restoration section 100includes a PLL section 104 for frequency acquisition, a PLL section 105for phase tracking, and a phase/frequency detector 101 used in commonfor the different PLL section 104 for frequency acquisition and PLLsection 105 for phase tracking.

[0068] Also, the carrier restoration section 100 includes a blinddecision element 102 and a decision-directed decision element 103 fordetermining the kind of decision signal constellations from an output ofthe PLL section for phase tracking, and operating the phase/frequencydetector 101 in a blind mode or decision-directed mode.

[0069] A lock detection section 14 determines an operation mode of thephase/frequency detector 101 of the carrier restoration section 100, andoutputs a corresponding control signal LD[2:0] to the phase/frequencydetector 101 of the carrier restoration section 100, a frequencyacquisition loop filter 104-1 of the PLL section 104 for frequencyacquisition, and a phase tracking loop filter 105-1 of the PLL section105 for phase tracking. The selection of an operation mode of thephase/frequency detector 101 is automatically performed by the lockcontrol signal LD[2:0] of the lock detection section 14.

[0070] Here, the PLL section 104 for frequency acquisition includes thefrequency acquisition loop filter 104-1 a numerically controlledoscillator (NCO) 104-2 and a frequency acquisition element 104-3. ThePLL section 105 for phase tracking includes the phase tracking loopfilter 105-1, a phase ROM table 105-2, and a phase tracking element105-3.

[0071] The phase/frequency detector 101 calculates the phase error intwo modes, i.e., a blind mode and a decision-directed mode, according tothe kind of decision signal constellations determined by the blinddecision element 102 and the decision-directed decision element 103.

[0072] The phase error generated from the phase/frequency detector 101is expressed as a polarity, and is outputted to the PLL section 104 forfrequency acquisition and the PLL section 105 for phase tracking.

[0073]FIG. 6 is a block diagram illustrating the detailed constructionof the phase/frequency detection section. The phase/frequency detectionsection 101 includes a first multiplexer 201 for selecting andoutputting one of an I blind decision signal D_(Blind) _(—) I outputtedfrom the blind decision element 102 and an I decision-directed decisionsignal D_(DD) _(—) I outputted from the decision-directed decisionelement 103 according to the control signal LD[1] generated from thelock detection section 14, a second multiplexer 202 for selecting andoutputting one of a Q blind decision signal D_(Blind) _(—) Q outputtedfrom the blind decision element 102 and a Q decision-directed decisionsignal D_(DD) _(—) Q outputted from the decision-directed decisionelement 103 a multiplier 203 for multiplying an output of the firstmultiplexer 201 by an I demodulated signal constellation CR_I generatedfrom the phase tracking element 105-3, a multiplier 204 for multiplyingan output of the second multiplexer 202 by a Q demodulated signalconstellation CR_Q generated from the phase tracking element 105-3, asubtracter 205 for calculating a difference between outputs of the twomultipliers 203 and 204 and outputting the phase error, and a polarityextraction section 206 for detecting a polarity of the phase errorPhase_Polarity outputted from the subtracter 205 in the unit of a symboland outputting the polarity of the phase error to the frequencyacquisition loop filter 104-1 and the phase tracking loop filter 105-1.

[0074]FIG. 7 is a block diagram illustrating the detailed constructionof the blind decision section. The blind decision section 102 includes apolarity extraction section 301 a for extracting a polarity of an Idemodulated signal constellation CR_I generated from the phase trackingsection 105-3 in the unit of a symbol, a third multiplexer 303 a forgenerating an I blind decision signal constellation D_(Blind) _(—) Iobtained by slicing by two levels the demodulated signal constellationaccording to the polarity extracted from the polarity extraction section301 a to output the I blind decision signal constellation D_(Blind) _(—)I to the phase/frequency detector 101, a polarity extraction section 301b for extracting a polarity of a Q demodulated signal constellation CR_Qgenerated from the phase tracking section 105-3 in the unit of a symbol,and a fourth multiplexer 303 b for generating a Q blind decision signalconstellation D_(Blind) _(—) Q obtained by slicing by two levels thedemodulated signal constellation according to the polarity extractedfrom the polarity extraction section 301 b to output the Q blinddecision signal constellation D_(Blind) _(—) Q to the phase/frequencydetector 101.

[0075] The blind decision element 102 generates the blind decisionsignal constellations D_(Blind) _(—) I and D_(Blind) _(—) Q obtained byslicing by two levels the demodulated decision signal constellationsCR_I and CR_Q according to the polarities of the demodulated signalconstellations CR_I and CR_Q generated from the phase tracking element105-3 irrespective of the level values of 4/16/64/256 QAM.

[0076]FIG. 8 is a view illustrating an example of the blind decisionsignal constellations of 4/16/64/256 QAM.

[0077]FIG. 9 is a block diagram illustrating the detailed constructionof the decision-directed decision element. The decision-directeddecision element 103 includes a multi-level comparator 401 a forcomparing levels of an I demodulated signal constellation CR_I generatedfrom the phase tracking element 105-3, a fifth multiplexer 403 a forselecting an I decision signal constellation D_(DD) _(—) I matching therespective signal levels of the I demodulated signal constellation CR_Iaccording to an output of the multi-level comparator 401 a to output theI decision signal constellation D_(DD) _(—) I to the phase/frequencydetector 101, a multi-level comparator 401 b for comparing levels of a Qdemodulated signal constellation CR_Q generated from the phase trackingelement 105-3, a sixth multiplexer 403 b for selecting a Q decisionsignal constellation D_(DD) _(—) Q matching the respective signal levelsof the Q demodulated signal constellation CR_Q according to an output ofthe multi-level comparator 401 b to output the Q decision signalconstellation D_(DD) _(—) Q to the phase/frequency detector 101.

[0078] The decision-directed decision element 103 generates the decisionsignal constellations D_(DD) _(—) I and D_(DD) _(—) Q matching therespective signal levels of the demodulated signal constellations CR_Iand CR_Q generated from the phase tracking element 105-3.

[0079]FIG. 10 is a view illustrating an example of constellations of the16 QAM decision-directed decision signal.

[0080]FIG. 11 is a block diagram illustrating the detailed constructionof the frequency acquisition loop filter. The frequency acquisition loopfilter 104-1 includes a seventh multiplexer 503 a for selecting oneamong a plurality of pre-calculated first positive bandwidth values 501a according to the control signal LD[2:0] of the lock detection section14, an eighth multiplexer 504 a for selecting one among a plurality ofpre-calculated first negative bandwidth values 502 a according to thecontrol signal LD[2:0] of the lock detection section 14, a ninthmultiplexer 505 a for selecting one of outputs of the seventh and eighthmultiplexers 503 a and 504 a according to the polarity of the phaseerror detected by the phase/frequency detector 101, a tenth multiplexer503 b for selecting one among a plurality of pre-calculated secondpositive bandwidth values 501 b according to the control signal LD[2:0]of the lock detection section 14, an eleventh multiplexer 504 b forselecting one among a plurality of pre-calculated second negativebandwidth values 502 b according to the control signal LD[2:0] of thelock detection section 14, a twelfth multiplexer 505 b for selecting oneof outputs of the tenth and eleventh multiplexers 503 b and 504 baccording to the polarity of the phase error detected by thephase/frequency detector 101, an adder 506 for adding an output of thetwelfth multiplexer 505 b and a feedback signal delayed by one symbol, adelay 507 for delaying an output of the adder 506 by one symbol andfeeding back the delayed output to the adder 506, an adder 508 foradding an output of the ninth multiplexer 505 a and an output of thedelay 507, and an adder 509 for generating a frequency offset by addingan output of the adder 508 and an intermediate frequency ω_(c) of thecarrier externally provided, and outputting the frequency offset to thenumerically controlled oscillator 104-2. Here, the adders 506 and 508,and the delay 507 comprise a kind of integrator, and generate thefrequency offset Δω by accumulating output results of the ninth andtwelfth multiplexers 505 a and 505 b in the unit of a symbol.

[0081] Specifically, in order to acquire the corresponding frequencyoffset Δω, the frequency acquisition loop filter 104-1 serves as a firstdigital low-pass filter for generating the corresponding frequencyoffset Δω by accumulating values of the positive or negative bandwidthsFrequency#BW_# according to the polarity of the phase error.

[0082] At this time, a gear shifting of the filter bandwidth isautomatically performed by the lock control signal LD[2:0] of the lockdetection section 14.

[0083]FIG. 12 is a block diagram illustrating the detailed constructionof the numerically controlled oscillator (NCO). The NCO 104-2 includes adelay 601 for delaying by one symbol a corresponding frequency offset(ω_(c)+Δω) outputted from the frequency acquisition loop filter 104-1 anadder 602 for adding an output of the delay 601 and a feedback signal, amodulo 2π 603 for calculating an output of the adder 602 by a 27 module,an adder 604 for delaying by one symbol an output of the modulo 2π 603and feeding back the delayed output to the adder 602, a cosine lookuptable 605 for storing a plurality of cosine waves, selecting andoutputting to the frequency acquisition element 104-3 a cosine wavecos(ω_(c)+Δω) corresponding to an output of the delay 604, and a sinelookup table 606 for storing a plurality of sine waves, selecting andoutputting to the frequency acquisition element 104-3 a sine wavesin(ω_(c)+Δω) corresponding to the output of the delay 604. Here, theadder 602, modulo 2π 603, and delay 604 comprise a simple integrator.

[0084] The NCO 104-2 generates the sine wave sin(ω_(c)+Δω) and thecosine wave cos(ω_(c)+Δω) in a digital form according to thecorresponding frequency offset (ω_(c)+Δω) generated from the frequencyacquisition loop filter 104-1.

[0085]FIG. 13 is a block diagram illustrating the detailed constructionof the frequency acquisition element. The frequency acquisition element104-3 includes a multiplier 701 for shifting an I base-band digitalsignal BB_I by multiplying the cosine wave cos(ω_(c)+Δω)) outputted fromthe NCO 104-2 and an I pass-band digital signal PB_Data outputted fromthe preprocessing section 11, and a multiplier 702 for shifting a Qbase-band digital signal BB_Q by multiplying the sine wave sin(ω_(c)+Δω)outputted from the NCO 104-2 and a Q pass-band digital signal PB_Dataoutputted from the preprocessing section 11.

[0086] Specifically, the frequency acquisition element 104-3 convertsthe pass-band digital signal PB_Data outputted from the preprocessingsection 11 where the frequency offset Δω is acquired by demodulating thepass-band digital signal PB_Data by the cosine wave cos(ω_(c)+Δω) andthe sine wave sin(ω_(c)+Δ) generated from the NCO 104-2.

[0087]FIG. 14 is a block diagram illustrating the detailed constructionof the phase tracking loop filter. The phase tracking loop filter 105-1includes thirteenth multiplexer 802 a for selecting one among aplurality of pre-calculated positive bandwidth values 801 a according tothe control signal LD[2:0] of the lock detection section 14, afourteenth multiplexer 802 b for selecting one among a plurality ofpre-calculated negative bandwidth values 801 b according to the controlsignal LD[2:0] of the lock detection section 14, a fifteenth multiplexer803 for selecting one of outputs of the thirteenth and fourteenthmultiplexers 802 a and 802 b according to the polarity of the phaseerror detected by the phase/frequency detector 101, an adder 804 foradding an output of the fifteenth multiplexer 803 and a feedback signal,a modulo π/4 805 for calculating an output of the adder 804 by a π/4module, and a delay 806 for delaying an output of the modulo π/4 805 byone symbol, feeding back the delayed output to the adder 804, andoutputting the delayed output to a phase ROM table 105-2. Here, theadder 804, modulo π/4 805, and delay 806 comprise a simple integrator.

[0088] Specifically, in order to track the corresponding phase jitterΔθ, the phase tracking loop filter 105-1 serves as a first digitallow-pass filter for generating the corresponding phase jitter Δθ byaccumulating values of bandwidth PhaseBw_# of the phase tracking loopfilter according to the polarity of the phase error.

[0089] At this time, a gear shifting of the filter bandwidth of thephase tracking loop filter 105-1 is automatically performed by the lockcontrol signal LD[2:0] of the lock detection section 14.

[0090]FIG. 15 is a block diagram of the detailed construction of thephase ROM table. The phase ROM table 105-2 includes an MSB extractionsection 905 for extracting only the most significant bit (MSB) of thephase jitter Δθ outputted from the phase tracking loop filter 105-1, alower bit extraction section 902 for extracting remaining bits exceptfor the MSB of the phase jitter Δθ outputted from the phase trackingloop filter 105-1, a 2's complement section 903 for obtaining acomplement on 2 with respect to an output of the lower bit extractionsection 902, a sixteenth multiplexer 904 from selecting one of an outputof the lower bit extraction section 902 and an output of the 2'scomplement section 903 according to an output of the MSB extractionsection 901, a lookup table 905 for selecting and outputting the sineand cosine waves corresponding to an output of the sixteenth multiplexer904, a 2's complement section 906 for obtaining a complement on 2 withrespect to the sine wave selected and outputted by the lookup table 905,and a seventeenth multiplexer 907 for selecting one of the sine waveoutputted from the lookup table 906 and the sine wave outputted from the2's complement section 903 according to the output of the MSB extractionsection 901.

[0091] That is, phase ROM table 105-2 generates the sine wave sin(Δθ)and the cosine wave cos(Δθ) according to the corresponding phase jitterΔθ generated from the phase tracking loop filter 105-1.

[0092]FIG. 16 is a block diagram illustrating the detailed constructionof the phase tracking element. The phase tracking element 105-3 includesa multiplier 911 for multiplying an I base-band digital signal BB_Ioutputted from the phase acquisition element 104-3 and the cosine wavecos(Δθ) outputted from the phase ROM table 105-2, a multiplier 912 formultiplying a Q base-band digital signal BB_Q outputted from the phaseacquisition element 104-3 and the sine wave sin(Δθ) outputted from thephase ROM table 105-2, an adder 915 for adding outputs of the twomultipliers 911 and 912 and outputting an I demodulated signalconstellation CR_I where the carrier is restored, a multiplier 913 formultiplying the I base-band digital signal BB_I outputted from the phaseacquisition element 104-3 and the sine wave sin(Δθ) outputted from thephase ROM table 105-2, a multiplier 914 for multiplying the Q base-banddigital signal BB_Q outputted from the phase acquisition element 104-3and the cosine wave cos(Δθ) outputted from the phase ROM table 105-2,and an adder 916 for obtaining subtraction of outputs of the twomultipliers 913 and 914 and outputting a Q demodulated signalconstellation CR_Q where the carrier is restored.

[0093] That is, the phase tracking element 1-5-3 tracks thecorresponding phase jitter Δθ of the base-band digital signals BB_I andBB_Q shifted in the frequency acquisition element 104-3 using the cosinewave cos (Δθ) and the sine wave sin(Δθ) generated from the phase ROMtable 105-2, and generates the base-band digital signals CR_I and CR_Qwhere the carrier is completely restored.

[0094] The acquisition/tracking performance of the carrier restorationsection according to the present invention is determined through analgorithm of the phase/frequency detector 101 and an implementationmethod of PLL.

[0095] Accordingly, the phase/frequency detector 101 of the carrierrestoration section 100 according to the present invention acquires thefrequency offset (Δθ) and tracks the residual phase jitter (Δθ) in twomodes. That is, the phase/frequency detector 101 performs a blind modefor acquiring the frequency offset (Δθ) and a decision-directed mode fortracking the residual phase jitter according to the kind of the useddecision signal constellations (I.e., outputs of the blind decisionelement 102 and decision-directed decision element 103 ). At this time,the mode selection of the phase/frequency detector 101 is automaticallyperformed by the lock detection section 14.

[0096] Specifically, two PLLs are provided in the carrier restorationsection 100. For example, the carrier restoration section 100 iscomposed of the PLL section 104 for frequency acquisition for acquiringthe frequency offset (Δθ) and the PLL section 105 for phase tracking fortracking the residual phase jitter (Δθ). At this time, thephase/frequency detector 101 is commonly used by the PLL section 104 forfrequency acquisition and the PLL section 105 for phase tracking.

[0097] Also, the phase/frequency detector 101 extracts the polarity byobtaining the phase error, and then expresses the phase error by thepolarity. This feature reduces the circuit complexity when implementingthe loop filter circuit.

[0098]FIG. 6 is a block diagram illustrating the detailed constructionof the phase/frequency detection section. The first and secondmultiplexers 201 and 202 selects and outputs to the multipliers 203 and204 one of the blind decision signal constellations D_(Blind) _(—) I andD_(Blind) _(—) Q generated from the blind decision element 102 and thedecision-directed decision signal constellations D_(DD) _(—) I andD_(DD) _(—) Q generated from the decision-directed decision element 103according to the control signal LD[1] generated from the lock detectionsection 14. The multiplier 203 multiplies the I demodulated signalconstellation CR_I outputted from the phase tracking element 105-3 bythe I demodulated signal constellation D_(Blind) _(—) I or D_(DD) _(—) Ito output the multiplied result to the subtracter 205. The multiplier204 multiplies the Q demodulated signal constellation CR_Q outputtedfrom the phase tracking element 105-3 by the Q demodulated signalconstellation D_(Blind) _(—) Q or D_(DD) _(—) Q to output the multipliedresult to the subtracter 205. The subtracter 205 calculates thedifference between the outputs of the two multipliers 203 and 204. As aresult, the output of the subtracter 205 will be the phase error betweenthe decision signal constellation and the demodulated signalconstellation.

[0099] The phase error obtained by the subtracter 205 is inputted to thephase extraction section 206, and the phase extraction section 206extracts only the polarity from the phase error. The extracted polarityof the phase error Phase_Polarity is outputted to the frequencyacquisition loop filter 104-1 of the PLL section 104 for frequencyacquisition and the phase tracking loop filter 105-1 of the PLL section105. The output of the phase/frequency detector 101 is one among {+1, 0,−1}.

[0100] At this time, the selection of the operation mode of thephase/frequency detector 101 is performed by the control signal LD[1 ]generated from the lock detection section 14.

[0101] That is, the first mode is for acquiring the frequency offset Δωof the carrier before an eye pattern of the demodulated signalconstellations CR_I and CR_Q opens due to the frequency offset Δω of thecarrier, and is called the blind mode. In the blind mode, if thefrequency offset Δω is acquired, the eye pattern of the demodulatedsignal constellations starts to open.

[0102] The second mode is for tracking the low frequency offset Δω andresidual phase jitter Δω of the carrier acquired through the blind mode,and is called the decision-directed mode.

[0103] If the characteristic function of the phase/frequency detector0101 is e(φ), it satisfies three conditions as shown in the followingequations 12 to 14, and the phase/frequency detector 101 of the M-QAMcarrier restoration section 100 can stably operate. e(φ)=e(φ+(½)*k*π)kεZ  [Equation 12]

e(φ)=−e(−φ)  [Equation 13]

If e(φ)=0, only φ=0 exists through [π/4, −π/4].  [Equation 14]

[0104] Here, φ is the difference between the phases of the demodulatedsignal constellation and the decision signal constellation, and Z is aninteger set.

[0105] Since the first condition of Equation 12 is that four quadrantsare not discriminated in case of the QAM, and it corresponds to thephase ambiguity of 90°. This phase ambiguity can be solved by performingthe differential encoding in the transmitting end, and performing thedifferential decoding in the receiving end.

[0106] The second condition of Equation 13 means the polarity of thephase difference between the demodulated signal constellation and thedecision signal constellation, which means that the phase error has apositive value or negative value according to the late/early state ofthe frequencies of the demodulated signal constellation and the localoscillator.

[0107] The third condition of Equation 14 means that the output of thephase/frequency detector 101 is 0 (i.e., zero) only when the phase ofthe demodulated signal constellation coincides with the phase of thedecision signal constellation.

[0108] Accordingly, the characteristic function e(φ) of thephase/frequency detector 101 is expressed by the following equations 15and 16 according to the two operation modes.

[0109] The characteristic function e(φ) of the phase/frequency detector101 in the blind mode is

e(φ)=sgn(θ−φ)=sgn(CR _(—) Q*D _(Blind) _(—) I−CR _(—) I*D _(Blind) _(—)Q)[Equation 15]  

[0110] The characteristic function e(φ) of the phase/frequency detector101 in the decision-directed mode is

e(φ)=sgn(θ−φ)=sgn(CR _(—) Q*D _(DD) _(—) I−CR _(—) I*D _(DD) _(—)Q)  [Equation 16]

[0111] Here, the sgn(#) operand serves as an extractor for extractingthe polarity #. Also, (CR_I, CR_Q) represent the inphase and thequadrature of the demodulated signal constellation, and θ represents thephase of the demodulated signal constellation. Also, D_(Blind) _(—) Iand D_(Blind) _(—) Q represent the inphase and the quadrature of theblind decision element 102 in the blind mode, and φ represents the phaseof the blind decision signal constellation. Especially, the phase φ ofthe decision signal constellation of the blind decision element 102 hasthe following values, and an example of the decision signalconstellation of 4/16/64/256 QAM is illustrated in FIG. 8.

[0112] First quadrant: φ=45°

[0113] Second quadrant: φ=135°

[0114] Third quadrant: φ=225°

[0115] Fourth quadrant: φ=315°

[0116] Also, α values of the respective quadrants are given in thefollowing table 1. α First Second Third Fourth Modulation QuadrantQuadrant Quadrant Quadrant 256-QAM 10.63 −10.63 −10.63 10.63 64-QAM 10.5−10.5 −10.5 10.5 16-QAM 10.0 −10.0 −10.0 10.0 4-QAM 8.0 −8.0 −8.0 8.0

[0117] An equation for calculating α values

α=(Σx ²)÷(Σabs(x))

[0118] where, x denotes the demodulated signal constellation.

[0119] Meanwhile, D_(DD) _(—) I and D_(DD) _(—) Q represent the inphaseand the quadrature of the decision-directed decision element 103 in thedecision-directed mode, and φ represents the decision signalconstallation in the decision-directed mode. FIG. 10 shows an example ofthe decision signal constellation of 16 QAM.

[0120]FIGS. 17A and 17B are views illustrating the geometricalcharacteristic of the characteristic function e(φ) of a phase/frequencydetector 101 in the blind mode. Specifically, FIG. 17A shows the casethat the phase θ of the demodulated signal constellations CR_I and CR_Qis larger than the phase φ of the decision signal constellationsD_(Blind) _(—) I and D_(Blind) _(—) Q, and the result of thecharacteristic function e(φ) of the phase/frequency detector 101 has apositive value (i.e., sgn(θ−φ)>0). FIG. 17B shows the case that thephase θ of the demodulated signal constellations CR_I and CR_Q issmaller than the phase φ of the decision signal constellations D_(Blind)_(—) I and D_(Blind) _(—) Q, and the result of the characteristicfunction e(φ) of the phase/frequency detector 101 has a negative value(i.e., sgn(θ−φ)<0).

[0121]FIGS. 18A and 18B are views illustrating the geometricalcharacteristic of the characteristic function e(φ) of a phase/frequencydetector 101 in the decision-directed mode. Specifically, FIG. 18A showsthe case that the phase θ of the demodulated signal constellations CR_Iand CR_Q is larger than the phase φ of the decision-directed decisionsignal constellations D_(DD) _(—) I and D_(DD) _(—) Q, and the result ofthe characteristic function e(φ) of the phase/frequency detector 101 hasa positive value (i.e., sgn(θ−φ)>0). FIG. 18B shows the case that thephase θ of the demodulated signal constellations CR_I and CR_Q issmaller than the phase φ of the decision-directed decision signalconstellations D_(DD) _(—) I and D_(DD) _(—) Q, and the result of thecharacteristic function e(φ) of the phase/frequency detector 101 has anegative value (i.e., sgn(θ−φ)<0).

[0122] Referring to the construction of the blind decision element 102as illustrated in FIG. 7, the polarity extraction sections 301 a and 301b extract the polarities of the demodulated signal constellations CR_Iand CR_Q generated from the phase tracking section 105-3, respectively,and provide the polarities to the third and fourth multiplexers 303 aand 303 b as selection signals. At this time, to the third and fourthmultiplexers 303 a and 303 b are inputted pre-calculated α values of therespective quadrants and inverted {overscore (α)} values 302 a and 302b, and one of the α value and the {overscore (α)} value is selected andoutputted according to the extracted polarity. That is, the outputs ofthe third and fourth multiplexers 303 a and 303 b become the 2-levelblind decision signal constellations D_(Blind) _(—) I and D_(Blind) _(—)Q.

[0123]FIG. 8 shows (I, Q) coordinates of the blind decision signalconstellations of 4/16/64/256 QAM. The blind decision signalconstellations generated from the third and fourth multiplexers 303 aand 303 b are used as the decision signal constellations when theoperation mode of the phase/frequency detector 101 is the blind mode.

[0124] Referring to the construction of the decision-directed decisionelement 103 the multi-level comparators 401 a and 401 b compare thesignal levels of the demodulated signal constellations CR_I and CR_Qgenerated from the phase tracking element 105-3, and provide the resultof comparison to the fifth and sixth multiplexers 403 a and 403 b,respectively. At this time, the n predetermined decision signal levelvalues 402 a and 402 b are inputted to the fifth and sixth multiplexers403 a and 403 b, and the fifth and sixth multiplexers 403 a and 403 bselect and output to the phase/frequency detector 101 one among the ndecision signal levels according to the output results of thecomparators 401 a and 401 b as the decision-directed decision signalconstellations D_(DD) _(—) I and D_(DD) _(—) Q. That is, thedecision-directed decision signal constellations D_(DD) _(—) I andD_(DD) _(—) Q outputted from the fifth and sixth multiplexers 403 a and403 b are used as the decision signal constellations when the operationmode of the phase/frequency detector 101 is the decision-directed mode.

[0125]FIG. 10 shows (I, Q) coordinates of the 4-level decision-directeddecision signal constellations of 16 QAM. For example, if thedemodulated signal constellations CR_I and CR_Q are within the decisionregion of the first quadrant, it is judged that they are the signals inthe first quadrant, and the decision-directed decision signalconstellation are generated accordingly.

[0126]FIG. 11 is a block diagram illustrating the detailed constructionof the frequency acquisition loop filter 104-2. The bandwidth values 501a, 501 b, 502 a and 502 b are previously calculated based on thefollowing Table 2, and are inputted to the seventh eighth, tenth andeleventh multiplexers 503 a, 503 b, 504 a, and 504 b. Specifically, thefirst positive bandwidth values Frequency1Bw_# are inputted to theseventh multiplexer 503 a, while the first negative bandwidth values(Frequency1Bw_#)-bar are inputted to the eighth multiplexer 504 a, basedon the table 2. The second positive bandwidth values Frequency2Bw_# areinputted to the tenth multiplexer 503 b, while the second negativebandwidth values (Rrequency2Bw_#)-bar are inputted to the eleventhmultiplexer 504 b, based on the table 2. TABLE 2 Word Dynamic Bandwidthof Floating Length Range Loop Filter Point Fixed Point 30Bits (0˜2 π) 2π 6.283185307 1073741823 Center 1.570796326 268435456 Frequency(π/2)Frequency1Bw_0 0.003972973 678912 Frequency1Bw_1 0.000529729 90496Frequency1Bw_2 0.000264865 45184 Frequency1Bw_3 0.000026486 4608Frequency2Bw_0 0.000080533 13824 Frequency2Bw_1 0.000014321 256Frequency2Bw_2 0.000003581 128 Frequency2Bw_3 0.000000006 1

[0127] Then, the seventh and eighth multiplexers 503 a and 504 a selectone among a plurality of the first positive bandwidth values and oneamong a plurality of the first negative bandwidth values, respectively,to output the selected values to the ninth multiplexer 505 a accordingto the control signal LD[2:0] of the lock detection section 14. Theninth multiplexer 505 a selects the first positive bandwidth values orthe first negative bandwidth values outputted from the seventh andeighth multiplexers 503 a and 504 a to output the selected values to theadder 508 according to the polarity of the phase error detected by thephase/frequency detector 101.

[0128] Also, the tenth and eleventh multiplexers 503 b and 504 b selectone among a plurality of the second positive bandwidth values and oneamong a plurality of the second negative bandwidth values, respectively,to output the selected values to the twelfth multiplexer 505 b accordingto the control signal LD[2:0] of the lock detection section 14. Thetwelfth multiplexer 505 b selects the second positive bandwidth valuesor the second negative bandwidth values outputted from the tenth andeleventh multiplexers 503 b and 504 b to output the selected values tothe adder 506 according to the polarity of the phase error detected bythe phase/frequency detector 101.

[0129] The adder 506 adds the output of the twelfth multiplexer 505 band the signal delayed by one symbol to output the result of addition tothe delay 507, and the delay 507 delays the output of the adder 506 byone symbol to output the delayed output to the adders 506 and 508. Theadder 508 adds the output of the ninth multiplexer 505 a and the outputof the delay 507 to output the result of addition to the adder 509. Theoutput of the adder 508 is the frequency offset Δω.

[0130] The adder 509 adds the frequency offset Δω outputted from theadder 508 and the intermediate frequency ω_(c) of the carrier externallyinputted to output the result of addition to the numerically controlledoscillator 104-2.

[0131] That is the adders 506 and 508, and the delay 507 comprise a kindof integrator, and generate the frequency offset Δω by accumulating theoutput results of the ninth and twelfth multiplexers 505 a and 505 b inthe unit of a symbol.

[0132]FIG. 12 is a block diagram illustrating the detailed constructionof the numerically controlled oscillator (NCO) 104-2. The NCO 104-2 is atypical numerically controlled oscillator for generating the digitaltype sine wave sin(ω_(c)+Δω) and the cosine wave cos(ω_(c)+Δω) accordingto the intermediate frequency ω_(c) and the frequency offset Δω of thecarrier wave generated from the frequency acquisition loop filter 104-1.In FIG. 12, the adder 602, the 2π module 603, and the delay 604 comprisea simple integrator, and use the phase characteristic value of themodulo 2π to prevent the overflow as known in the art. The sine wavesin(ω_(c)+Δω) and the cosine wave cos(ω_(c)+Δω) corresponding to thesignal outputted from the integrator are selected from the cosine lookuptable 605 storing a plurality of cosine waves and the sine lookup table606 storing a plurality of sine waves, and are outputted to thefrequency acquisition element 104-3.

[0133]FIG. 13 is a block diagram illustrating the detailed constructionof the frequency acquisition element. The multiplier 701 multiplies thecosine wave cos(ω_(c)+Δω) outputted from the NCO 104-2 and the Ipass-band digital signal PB_I outputted from the preprocessing section11 to shift the I pass-band digital signal PB_I to the I base-banddigital signal BB_I. The multiplier 702 multiplies the sine wavesin(ω_(c)+Δω) outputted from the NCO 104-2 and a Q pass-band digitalsignal PB_Q outputted from the preprocessing section 11 to shift the Qpass-band digital signal PB_Q to the Q base-band digital signal BB_Q.

[0134] Specifically, the frequency acquisition element 104-3 demodulatesthe pass-band digital signal PB_Data having the frequency offset Δωgenerated from the preprocessing section 11 by the cosine wavecos(ω_(c)+Δω) and the sine wave sin(ω_(c)+Δω) generated from the NCO104-2 and outputs the base-band digital signals BB_I and BB_Q with thefrequency offset Δω acquired, i.e., compensated for.

[0135]FIG. 14 is a block diagram illustrating the detailed constructionof the phase tracking loop filter 105-1, in which bandwidth values 801 aand 801 b of the phase tracking loop filter are pre-calculated based onthe following table 3, and are inputted to the thirteenth and fourteenthmultiplexers 802 a and 802 b. Specifically, the positive bandwidthvalues PhaseBw_# is inputted to the thirteenth multiplexer 802 a, whilethe negative bandwidth values (PhaseBw_#)-bar is inputted to thefourteenth multiplexer 802 b, based on the table 3. TABLE 3 Word DynamicBandwidth of Floating Length Range Loop Filter Point Fixed Point 20Bits(−π/4˜π/4) π/4 0.785398164 524288 PhaseBw_0 0.057268079 38228 PhaseBw_10.000572681 382 PhaseBw_2 0.000143175 95 PhaseBw_3 0.000001432 1

[0136] The thirteenth and fourteenth multiplexers 802 a and 802 b selectone among a plurality of the positive bandwidth values and one among aplurality of the negative bandwidth values, respectively, to output theselected values to the fifteenth multiplexer 803 according to thecontrol signal LD[2:0] of the lock detection section 14. The fifteenthmultiplexer 803 selects the positive or negative bandwidth valuesoutputted from the thirteenth and fourteenth multiplexers 802 a and 802b to output the selected value to the adder 804 according to thepolarity of the phase error detected by the phase/frequency detector101.

[0137] The output of the adder 804 is successively fed back to the adder804 through the modulo π/4 805 and the delay 806, and simultaneously isoutputted to the phase ROM table 105-2. Specifically, the adder 804 addsthe output of the fifteenth multiplexer 803 and the feedback signal tooutput the result of addition to the modulo π/4 805. Here, the adder804, the π/4 module 805, and the delay 806 comprise a simple integrator.

[0138] Specifically, the integrator generates the residual phase jitterΔθ of the carrier wave by accumulating the output result of thefifteenth multiplexer 803 in a unit of symbol. The generated residualphase jitter Δθ of the carrier is inputted to the phase ROM table 105-2.

[0139]FIG. 15 is a block diagram of the detailed construction of thephase ROM table. The phase ROM table 105-2 generates the sine wavesin(ω_(c)+Δω) and the cosine wave cos(ω_(c)+Δω) in the range of −π/4˜π/4according to the residual phase jitter Δθ of the carrier generated fromthe frequency acquisition loop filter 105-1 to output the sine andcosine waves to the phase tracking element 105-3.

[0140] The MSB extraction section 905 extracts the most significant bit(MSB), i.e., sign bit of the residual phase jitter Δθ of the inputtedcarrier, and provides the sign bit as a selection signal of thesixteenth and seventeenth multiplexers 904 and 905. The lower bitextraction section 902 extracts the remaining bits from the residualphase jitter Δθ of the carrier except for the MSB of the phase jitterΔθ. The output of the lower bit extraction section 902 is bypassed tothe sixteenth multiplexer 904, and simultaneously the 2's complementsection 903 obtains a complement on 2 with respect to an output of thelower bit extraction section 902 to output the 2's complement to thesixteenth multiplexer 904. The sixteenth multiplexer 904 selects one ofthe output of the lower bit extraction section 902 and the output of the2's complement section 903 to output the selected output to the lookuptable 905 according to the output of the MSB extraction section 901. Thelookup table 905 selects and outputs the sine and cosine wavescorresponding to the output of the sixteenth multiplexer 904. That is,the cosine wave cos(Δθ) is directly inputted to the phase tracker 105-3,and the sine wave sin(Δθ) is inputted to the phase tracking element105-3 via the seventeenth multiplexer 707.

[0141] The seventeenth multiplexer 907 selects one of the sine wave sin(Δθ) bypassed from the lookup table 906 and the sine wave obtaining the2's complement from the 2's complement section 903 to output theselected sine wave to the phase tracking element according to theMSBoutputted from the MSB extraction section 901.

[0142]FIG. 16 is a block diagram illustrating the detailed constructionof a phase tracking element 105-3. The phase tracking element 105-3demodulates the bass-band digital signal having the frequency offset Δωacquired by the frequency acquisition element 104-3 by the sine wavesin(Δθ) and the cosine wave cos(Δθ) in the range of −π/4˜π/4 producedfrom the phase ROM table 105-2, and generates the base-band digitalsignals CR_I and CR_Q with the residual phase jitter Δθ tracked by thephase tracking element 105-3. The base-band digital signals CR_I andCR_Q with the residual phase jitter Δθ tracked by the phase trackingelement 105-3 are outputted to the post-proceeding section 13, andsimultaneously are outputted to the blind decision element 102, thedecision-directed decision element 103 and the phase/frequency detector101.

[0143] As described above, the carrier restoration apparatus accordingto the present invention can be applied to all of QAM/PSK digitalreceivers.

[0144] For example, the carrier restoration apparatus can be applied toa single QAM cable digital receiver, a single QPSK satellite digitalreceiver, a single 8PSK satellite digital receiver, a composite QAM/QPSKcable/satellite digital receiver, a composite QAM/8PSK cable/satellitedigital receiver or the like.

[0145] With the construction of the carrier restoration apparatusaccording to the present invention, the frequency acquisition PLLsection for acquiring the frequency offset and the phase tracking PLLsection for tracking the residual phase jitter are separatelyconstructed, and the apparatus operates in two modes for first acquiringthe frequency offset and then tracking the residual phase jitter, sothat the rapid acquisition/tracking can be performed so as to minimizethe frequency offset and phase jitter of several hundred KHz producedfrom the tuner or the RF oscillator, and the high-reliabilityacquisition/tracking can be performed even under the low SNR and seriouschannel ISI (i.e., ghost).

[0146] Further, since the phase/frequency detector for detecting thephase error is commonly used for the frequency acquisition PLL sectionand the phase tracking PLL section, and the phase error is expressed bythe polarity, the circuit complexity can be reduced, and especially thecircuit construction of the frequency acquisition PLL section and thephase tracking PLL section can be simplified.

[0147] The forgoing embodiments are merely exemplary and are not to beconstrued as limiting the present invention. The present teachings canbe readily applied to other types of apparatuses. The description of thepresent invention is intended to be illustrative, and not to limit thescope of the claims. Many alternatives, modifications, and variationswill be apparent to those skilled in the art.

What is claimed is:
 1. A carrier restoration apparatus for converting apass-band digital signal of a specified channel into a base-band digitalsignal where a frequency offset and a phase jitter are restored bydemodulating the pass-band digital signal by a sine/cosine wave, theapparatus comprising: a phase/frequency detection section for obtaininga phase error between constellations of a demodulated signal andconstellations of a blind decision signal or a decision-directeddecision signal, and extracting a polarity of the phase error; a loopsection for frequency acquisition for extracting the correspondingfrequency offset by accumulating pre-calculated bandwidth valuesaccording to the polarity of the phase error, generating the digitaltype sine and cosine waves according to the extracted frequency offset,and then generating the base-band digital signal where the frequencyoffset of a carrier is acquired by demodulating the pass-band digitalsignal by the sine and cosine waves; a loop section for phase trackingfor extracting the corresponding phase jitter by accumulating thepre-calculated bandwidth values according to the polarity of the phaseerror, generating the digital type sine and cosine waves according tothe extracted phase jitter, and then generating demodulated signalconstellations where the phase jitter is tracked by demodulating thebase-band digital signal by the sine and cosine waves; a blind decisionsection for extracting the polarity of the demodulated signalconstellations generated from the loop section for phase tracking, andgenerating blind decision signal constellations by slicing thedemodulated signal constellations according to the extracted polarity;and a decision-directed decision section for generating adecision-directed decision signal constellations matching respectivesignal levels of the demodulated signal constellations generated fromthe loop section for phase tracking.
 2. The apparatus as claimed inclaim 1, wherein the phase/frequency detection section operates in ablind mode for extracting the polarity by obtaining the phase errorbetween the demodulated signal constellation and the blind decisionsignal constellation in order to acquire the frequency offset, or in adecision-directed mode for extracting the polarity by obtaining thephase error between the demodulated signal constellation and thedecision-directed decision signal constellation in order to track thephase jitter, and further comprises a lock detection section forcontrolling selection of the blind mode and the decision-directed mode.3. The apparatus as claimed in claim 1, wherein the loop section forfrequency acquisition comprises: a frequency acquisition loop filter fordetecting the corresponding frequency offset Δω by accumulating valuesof predetermined positive or negative bandwidths for frequencyacquisition according to the polarity of the phase error extracted bythe phase/frequency detection section; a controlled oscillator forgenerating the digital type sine wave sin(ω_(c)+Δω) and cosine wavecos(ω_(c)+Δω) according to the corresponding frequency offset detectedby the frequency acquisition loop filter; and a frequency acquisitionelement for generating the base-band digital signal where the frequencyoffset is acquired by demodulating the pass-band digital signal PB_Databy the cosine wave cos(ω_(c)+Δω) and the sine wave sin(ω_(c)+Δω)generated from the controlled oscillator.
 4. The apparatus as claimed inclaim 3, wherein a gear shifting of the filter bandwidth of thefrequency acquisition loop filter is automatically performed by a lockcontrol signal of a lock detection section.
 5. The apparatus as claimedin claim 3, wherein the frequency acquisition loop filter comprises: afirst selection section for selecting one among a plurality ofpre-calculated first positive bandwidth values according to a lockcontrol signal of a lock detection section; a second selection sectionfor selecting one among a plurality of pre-calculated first negativebandwidth values according to the lock control signal: a third selectionsection for selecting one of outputs of the first and second selectionsections according to the polarity of the phase error detected by thephase/frequency detector; a fourth selection section for selecting oneamong a plurality of pre-calculated second positive bandwidth valuesaccording to the lock control signal; a fifth selection section forselecting one among a plurality of pre-calculated second negativebandwidth values according to the lock control signal; a sixth selectionsection for selecting one of outputs of the fourth and fifth selectionsections according to the polarity of the phase error detected by thephase/frequency detector; and an integrator for detecting thecorresponding frequency offset Δω) by accumulating outputs of the thirdand sixth selection sections in the unit of a symbol.
 6. The apparatusas claimed in claim 1, wherein the loop section for phase trackingcomprises: a phase tracking loop filter for generating the correspondingphase jitter Δθ by accumulating values of pre-calculated bandwidths forphase tracking according to the polarity of the phase error extracted bythe phase/frequency detection section; a phase ROM table for generatingthe digital type sine wave sin(ω_(c)+Δω) and cosine wave cos(ω_(c)+Δω)according to the corresponding phase jitter detected by the phasetracking loop filter; and a phase tracking element for generating thedemodulated signal constellations where the corresponding phase jitterof the base-band digital signal outputted from the loop section forfrequency acquisition is tracked by the cosine wave cos(Δθ) and the sinewave sin(Δθ) generated from the phase ROM table.
 7. The apparatus asclaimed in claim 6, wherein a gear shifting of the filter bandwidth ofthe phase tracking loop filter is automatically performed by a lockcontrol signal of a lock detection section.
 8. The apparatus as claimedin claim 6, wherein the phase tracking loop filter comprises: a firstselection section for selecting one among a plurality of pre-calculatedfirst positive bandwidth values for phase tracking according to a lockcontrol signal of a lock detection section; a second selection sectionfor selecting one among a plurality of pre-calculated first negativebandwidth values for phase tracking according to the lock controlsignal; a third selection section for selecting one of outputs of thefirst and second selection sections according to the polarity of thephase error detected by the phase/frequency detector; and an integratorfor detecting the corresponding phase jitter Δθ by accumulating anoutput of the third selection sections in the unit of a symbol.
 9. Theapparatus as claimed in claim 1, wherein the blind decision sectioncomprises: a polarity extraction section for extracting the polarity ofthe demodulated signal constellation generated from the phase trackingsection in the unit of a symbol; and a selection section for generatingblind decision signal constellations by slicing by two levels thedemodulated signal constellations according to the polarity extracted bythe polarity extraction section; wherein pre-calculated a values andinverted {overscore (α)} values of respective quadrants are inputted tothe selection section, and the polarity from the polarity extractionsection is used as a selection signal.
 10. The apparatus as claimed inclaim 1, wherein the decision-directed decision section comprises: amulti-level comparator for comparing levels of the demodulated signalconstellations generated from the phase tracking element; and aselection section for receiving a plurality fo level signals, selectingone among the plurality of level signals using an output of themulti-level comparator as a selection signal, and outputting theselected level signal to the phase/frequency detection section as thedecision-directed decision signal constellations.
 11. A carrierrestoration method of converting a pass-band digital signal of aspecified channel into a base-band digital signal where a frequencyoffset and a phase jitter are restored by demodulating the pass-banddigital signal by a sine/cosine wave, the method comprising: aphase/frequency detection step of obtaining a phase error betweenconstellations of a demodulated signal and constellations of a blinddecision signal or a decision-directed decision signal, and extracting apolarity of the phase error; a frequency acquisition step of extractingthe corresponding frequency offset by accumulating pre-calculatedbandwidth values according to the polarity of the phase error,generating the digital type sine and cosine waves according to theextracted frequency offset, and then generating the base-band digitalsignal where the frequency offset of a carrier is acquired bydemodulating the pass-band digital signal by the sine and cosine waves;a phase tracking step of extracting the corresponding phase jitter byaccumulating the pre-calculated bandwidth values according to thepolarity of the phase error, generating the digital type sine and cosinewaves according to the extracted phase jitter, and then generatingdemodulated signal constellations where the phase jitter is tracked bydemodulating the base-band digital signal by the sine and cosine waves;a blind decision step of extracting the polarity of the demodulatedsignal constellations generated from the loop section for phasetracking, and generating blind decision signal constellations by slicingthe demodulated signal constellations according to the extractedpolarity; and a decision-directed decision step of generating adecision-directed decision signal constellations matching respectivesignal levels of the demodulated signal constellations generated at thephase tracking step.
 12. The method as claimed in claim 11, wherein thephase/frequency detection step extracts the polarity by obtaining thephase error between the demodulated signal constellation and the blinddecision signal constellation in a blind mode for acquiring thefrequency offset.
 13. The method as claimed in claim 11 wherein thephase/frequency detection step extracts the polarity by obtaining thephase error between the demodulated signal constellation and thedecision-directed decision signal constellation in a decision-directedmode for tracking the phase jitter.
 14. The method as claimed in claim11, wherein the phase/frequency detection step further comprises a lockdetection step for controlling selection of a blind mode and adecision-directed mode.
 15. The method as claimed in claim 14, whereinthe lock detection step automatically performs a gear shifting withrespect to respective filter bandwidth at the frequency acquisition stepand the phase tracking step.
 16. The method as claimed in claim 11,wherein the frequency acquisition step comprises the steps of: detectingthe corresponding frequency offset Δω by accumulating values ofpredetermined positive or negative bandwidths for frequency acquisitionaccording to the polarity of the phase error extracted at thephase/frequency detection step; generating the digital type sine wavesin(ω_(c)+Δω) and cosine wave cos(ω_(c)+Δω) according to thecorresponding frequency offset detected at the frequency offsetdetection step; and generating the base-band digital signal where thefrequency offset is acquired by demodulating the pass-band digitalsignal PB_Data by the cosine wave cos(ω_(c)+Δω) and the sine wavesin(ω_(c)+Δω) generated at the sine and cosine wave generating step. 17.The method as claimed in claim 1, wherein the phase tracking stepcomprises the steps of: generating the corresponding phase jitter Δθ byaccumulating values of pre-calculated bandwidths for phase trackingaccording to the polarity of the phase error extracted at thephase/frequency detection step; generating the digital type sine wavesin(ω_(c)+Δω) and cosine wave cos(ω_(c)+Δω) according to thecorresponding phase jitter detected at the phase jitter generating step;and generating the demodulated signal constellations where thecorresponding phase jitter of the base-band digital signal outputted atthe frequency acquisition step is tracked by the cosine wave cos(Δθ) andthe sine wave sin(Δθ) generated at the sine and cosine wave generatingstep.
 18. The method as claimed in claim 11, wherein the blind decisionstep comprises the steps of: extracting the polarity of the demodulatedsignal constellation generated at the phase tracking step in the unit ofa symbol; and generating blind decision signal constellations by slicingby two levels the demodulated signal constellations according to thepolarity extracted at the polarity extraction step.
 19. The method asclaimed in claim 11, wherein the decision-directed decision stepcomprises the steps of: comparing levels of the demodulated signalconstellations generated at the phase tracking step; and selecting oneamong a plurality of level signals according to a result of comparison,and outputting the selected level signal as the decision-directeddecision signal constellations of the phase/frequency step.